[04-FEB-21] The Animal Location Tracker (ALT) is a platform that we insert beneath an animal cage to track the movements of, and record telemetry signals from, subcutaneous transmitters (SCTs) and implantable stimulators (ISTs). The ALT provides an array of detector coils that both decode telemetry messages and measure their power. The power measurements made by the coils allow us to estimate the position of the transmitter, and to make reliable measurements of its location and velocity. Because each transmitter has its own unique channel number, we have no difficulty distinguishing between the animals in the cage. If we combine the ALT with synchronous video, such as recorded by our animal cage cameras (AACs), the ALT allows us to identify animals seen in the video by correlating their movements with the movements of their implanted transmitters.
The A3038A provides an array of fifteen coils on a 12-cm grid. The platform is 26 cm × 50 cm and is designed to provide reliable measurement of movement in a cage measuring 22 cm × 46 cm at the base. The ALT receives power and communication through a single Power over Ethernet (PoE) socket at the right side of the platform. To obtain reliable reception and movement measurement, the ALT must operate within a Faraday enclosure, such as our bench-top, stackable FE3A, or our canopy enclosure FE5A. We connect the ALT to an RJ-45 feedthrough in the wall of the enclosure with a shielded CAT-5 cable, and from there to a PoE switch with either a shielded or unshielded CAT-5 cable. We connect our data acquisition computer to the same PoE switch and so record signal and position data from the ALT to disk with our LWDAQ Software. Comparing the ALT to the components of our traditional telemetry system, the ALT combines the functions of our LWDAQ Driver (A2071E), Octal Data Receiver (A3027E), and Loop Antennas (A3015C) into one assembly with only one cable.
Our implanted transmitters emit 7-μs bursts of electromagnetic radiation in the range 902-928 MHz. Each burst contains a digital message. Each detector coil provides a measurement of radio-frequency power in this same frequency range. When one of the detector coils reports that a message transmission is in progress, all fifteen detector coils record the power they are receiving. The A3038 saves the transmitted message, along with fifteen eight-bit power values, in its memory, available for download by the Recorder Instrument in the LWDAQ Software, or download and storage to disk by the Neuroarchiver Tool. The Neuroarchiver contains a Tracker button that opens the Neurotracker Panel, which displays the measured position of transmitters on the tracker platform.
The A3038 is an improved version of our original A3032. The A3032 provided fifteen detector coils on a 32 cm × 16 cm grid, and used the coils to measure transmitter position, but did not itself decode the radio-frequency transmissions. The A3038 uses each detector coil both as a power meter and a data receiver. We present the basis of the ALT position measurement in the A3032 Feasibility Study and Development pages. We concluded that absolute accuracy of the ALT's position measurement for a transmitter in a beaker of water with respect to the coordinate system defined by its coil centers is ±25% of the coil pitch 90% of the time, and ±100% of the coil pitch the rest of the time. Because there is no way to know whether the current absolute position is accurate or not, the absolute position measurement is not useful. Its measurement of the direction and magnitude of movement, however, provide a reliable measurement of animal activity, and a reliable way to identify animals in video. The correlation between the ALT measurement of movement and video blob-tracking permits us to be 100% certain which blob corresponds to which animal, even when there are a dozen animals in the field of view.
The following versions of the A3038 and its sub-assemblies exist or are planned.
|Version||X (cm)||Y (cm)||Coil Pitch (cm)||Detector
|A3038A||51||27||12||A3038DM-B||15||Animal location tracker platform for 48 cm × 24 cm cage.|
|A3038DM-X||18||5||NA||33 nH SMT||1||Power and signal detector test circuit.|
|A3038DM-A||5||3||NA||250 nH SMT||1||Detector module prototype.|
|A3038DM-B||5||3||NA||250 nH SMT||1||Detector module used in A3038A.|
|A3038BB-X||51||27||12||MPCIE||15||Base board test fixture with sockets and power.|
|A3038BB-A||51||27||12||MPCIE||15||Base board assembly with logic, sockets, and power.|
[02-FEB-21] Current consumption fo the A3038DM-A detector module is 40 mA from 3.3 V. With fifteen of them deployed in the A3038A, plus control logic, embedded computer, and converters, we expect total current consumption from 3.3 V to be around 1.0 A. Current consumption from the 48-V PoE supply will be around 100 mA.
[06-APR-21] We program and calibrate the ALT Detector Module (A3028DM) with the Detector Module Programming and Calibration Station. The detector module mounts upon a circuit board that provides a socket for the module and a programming connector, as well as a 3.3 V power input. We program and calibrate before we load the antenna coild L1.
We drive the VCO with a 10-kHz 2-Vpp positive ramp, and trigger on the negative edge at the end of the ramp. We apply the −40 dBm sweep produced by the VCO and attenuators to A as shown above and look at P. We should see something like the trace below.
We should see the SAW filter pass band 900-930 MHz on P. We set up cursors to mark the two extremes of the pass band. Now we apply the sweep to B with UMC socket P2 and look at the demodulator output, D. We want the peak response of the tuner to be in the range 930-940 MHz, with 930 MHz being ideal. The ideal response is shown below.
We adjust the peak frequency by adding capacitors to C15. When we first assemble the board, we have C15 = 2.7 pF, which is certainly too low for peak frequency 930 MHz. Each 0.1 pF we add to C15 lowers the peak frequency by 10 MHz. We measure the peak frequency and add 0.1 pF and 0.2 pF capacitors so as to bring the peak as close to 930 MHz without going below 930 MHz. We must wash and dry the board after we add the capacitors, because flux residue will flatten the resonance and shift its frequency downwards by up to 10 MHz. For example final component values and peak frequencies see here.
Once the tuner is calibrated, we add another 10-dB attenautor to our sweep and apply the −50 dBm sweep to A with P1. We examine D and should see some adequate response like the one shown below.
Now that the detector is programmed and calibrated, we turn on our −8 dBm SCT signal generator and add 40 dB of attenuators to create a −48 dBm SCT signal. This we apply to A with P1 and check the blue LED to see if we have reception of the SCT signal. Reception should be robust. The blue light may flicker off once or twice a second, but will be on most of the time. Having passed these tests the detector module is ready for inclusion in an animal location tracker, where it will be further tested with moving transmitters in an enclosure.
[19-SEP-20] The A3038X requires the addition of four 10 nF and four 47 pF decoupling capacitors for U1-U4, as well as two 100 pF capacitors around U6.
[17-FEB-21] The A303801B printed circuit board needs the following corrections.
[14-JUL-20] We examine the response of the A3032C ALT amplifier and detector. We apply a −6 dBm sweep 840-980 MHz to a Loop Antenna (A3015C). We generate the sweep with a Modulating Transmitter (A3014MT), and we split the sweep and mix with 910 MHz to produce an IF ±70 MHz, which we run through a 21-MHz low-pass filter before viewing on the scope. We hold the loop antenna above Coil 10 on V0384 and observer the following response on PW, which is U1004-3.
We repeat on all fifteen coils and find the same response on each one, ±100 mV variation in 890-930 MHz, which is roughly ±3 dB. We remove the A3051C loop antenna and replace it with a 3-dB attenuator and a 50-mm bent wire antenna. We observe the same ±3 dB variation on power through the pass band. But in rare orientations of the antenna, all surrounding objects remaining stationary, received power drops suddenly, and variation is ±6 dB.
[29-JUL-20] We are considering using detector diode such as the SMS7630 in our coil amplifier to provide power limiting and power detection.
In SkyWorks application note APN1014, we see the detector circuit they used to obtain the above rectified voltage versus input power. They deliver power from a 50-Ω source, but do not load the source with 50 Ω. Instead, they place a diode and balast capacitor in series with the 50-Ω source impedance. The voltage across the diode will be roughly double what we would see if we loaded the source with 50 Ω. Their "Incident Power (dBm)" is the power that reflects off the detector circuit, which would be equal to the power delivered to a 50 Ω load. The "video resistance" they refer to in the detector plot is the resistor loading the balast capacitor.
We assemble a power detector made out of an SMS7630 diode and a 100 pF balast capacitor attached to a 50-Ω transmission line carrying the output of an A3029B amplifier. We measure the voltage across the balast capacitor, which we call the rectified voltage. We vary the transmission line power from −30 dBm to +28 dBm. Below −30 dBm our rectified voltage is swamped by noise. At +28 dBm, our amplifier is saturating. When we remove the diode and capacitor, our amplifier saturates at +30 dBm. With +6 dBm we add 1 kΩ in series with the diode and see only 90 mV, compared to 360 mV with no resistor.
We solder 51 Ω from the center pin of a BNC socket to ground. In parallel we place a SMS7630 diode and a 1.0 nF balast capacitor. We supply power down a 1-m coaxial cable from our synthesizer, and vary power from −30 dBm to +10 dBm. We measure the voltage rectified voltage versus power and plot.
[30-JUL-20] We solder a 33 kΩ resistor across a BNC socket. In parallel we place a SMS7630 diode and a 1.0 nF balast capacitor. We connect the detector directly to the output of our synthesizer. Without the 33 kΩ, we see no sustained rectified voltage, because the incoming power is capacitively coupled.
[07-AUG-20] We have the schematic of a prototype detector module, which provides both power measurement and demodulation of SCT messages, S3038X_1. Top view of circuit board here. This circuit will provide the D input for an A3007D so as to provide SCT signal reception, and will also produce an output P from the power meter. Power supply will come from a two-pin molex plug.
[04-SEP-20] We assemble a prototype detector module (A3038DM-X), as in S3038X, except we omit U5. We remove L1 so that we can supply A through J1. We remove R9 so that we extract B through J2. We connect 3.0V to P1. We see 33 mA flowing in. With four of BGA2803 and one of LT5534 we expect 31 mA. We connect J2 to our hand-held spectrum analyzer, and J1 to our frequency synthesizer, with attenuators as needed, to obtain the following plot of B versus A for 910 MHz.
The BGA2803 gain at 910 MHz is 24 dB. We have three 3-dB attenuators for −9 dB. We have the insertion loss of the SAQ filter L2, a B3588, which is around 2 dB. We expect the gain of the first two stages to be 37 dB. We see 41 dB from −90 to −50 dBm input. The output of U2 saturates at −5 dBm, which is consistent with the BG2803 data sheet.
We remove R1. Now we are shorting the input to ground with R17. We see 0.5 V at the output of the power detector, P, indicating −49 dBm at the input of the power detector, which implies −90 dBm at the input of U1, which is consistent with 50-Ω thermal noise in 900-930 MHz. When we restore R1, with our frequency synthesizer still attached, we see 1.8 V on P, meaning B is −17 dBm, which suggests interference of −58 dBm.
We apply −60 dBm to A and measure B as we vary frequency. We see the pass-band of the SAW filter clearly. We are surprised by the higher gain at 900 MHz and 930 MHz, which mark the edges of the pass-band. We restore R2 and sweep frequency again, this time measuring power at J4.
Output J4 is flat to within ±1 dB in the SAW filter pass band, at around −6 dBm. Diode arrays DA1 and DA2 are SMS7630. Their saturation current is 5 μA, so we expect their dynamic resistance to be 50 Ω for forward voltage 110 mV. Our guess was they would limit the amplitude to −3 dBm. But B and J4 are limited to −5 dBm by the BGA2803 saturation alone. We predict that removing DA1 and DA2 will change nothing. We would like B to lie in the range −60 dBm to 0 dBm when detecting transmitter power bursts. But we can receive up to −20 dBm from a transmitter held close to a detector coil. The gain from A to B is 21 dB too high.
[07-SEP-20] We apply −52 dBm 910 MHz to A. We have L1 and R9 removed. We measure −10 dBm at B. Gain is 42 dBm. We mix with +7dBm of 880 MHz using a ZAD-11 and see IF amplitude is 50 mVrms, or −13 dBm (conversion loss only −5 dBm). We sweep the frequency from 820-980 MHz and obtain the following IF trace.
Gain in this measurement is constant to ±2 dB in the pass-band of the SAW filter. We do not see the +10 dB ears that appear in our measurement using our hand-held spectrometer. For −10 dBm we expect 2.2 V at P. We see 2.4 V. We vary power supply voltage and measure gain and supply current.
We restore R9 and apply −36 dB sweep 820-980 MHz to A. We mix the same sweep with 910 MHz to produce IF that we pass through 21 MHz low-pass filter. We have L3 = 3.3 nH, C16 = 0.5 pF, C17 = 2.0 pF, C18 = 10 pF, R25 omitted. We have U5 loaded and feedback configured for D = 4R. We look see the following tuning curve on D.
We see D go from 0.5 V at 894 MHz to 2.5 V at 936 MHz, which is perfect for demodulating SCT signals. We apply an SCT signal −31 dB, to A. With no R25 and C18 = 10 pF, our SCT demodulated signal is a triangle wave. The rise time of D appears to be around 200 ns. Add R25 = 1 kΩ and see the trace below, rise time around 50 ns.
The above signal is similar to the traces we see in our downshifting receivers. We repeat our frequency sweep to see again the tuning response. We now have D going from 0.2 to 0.9 V in the band 894-936 MHz.
We try C18 = 0 pF with R25 omitted. We see the same demodulation sweep amplitude as for C18 = 10 pF with R25 omitted, but when we apply an SCT signal, the rise time is around 100 ns.
[10-SEP-20] We remove R1. We connect C13 to R6/R7 instead of B. We see 0.74 V on P. We load 100 pF from U6-4 to 0V and see 0.40 V. Adding another 100 pF makes no difference, nor does adding 10 pF. We load 100 pF across P1 and see 0.35 V on P. As soon as we restore R1, even without L1, we see 1.1 V. If we connect 50 Ω to J1 we see 0.44 V. With L1 loaded and J1 open circuit 1.1 V. Restore C13 to its previous connection and see 2.1 V on P. Remove R1 and see 1.2 V.
[12-SEP-20] Restore R1 and see 2.1 V on P. We test the hypothesis that J1 and L1 are picking up radio waves transmitted by J4, and the 80-dB total gain of the amplifier is generating oscillations at a frequency above 2.4 GHz. We rotate C7 and C10 so that they connect U3-6 and U4-6 to 0V. We still see 2.1 V on P. We rotate C4 to connect U2-6 to 0V. We connect C13 to R7, so we have U2, U3, U4 with grounded inputs, their outputs loaded by attenuators, and U6-6 driven by R7. With L1 loaded and J1 open circuit, P = 1.1 V. Remove L1, P = 1.1 V. J1 is not picking up radio waves from J4.
[14-SEP-20] We restore R1, restore C13 to B, and restore C4. Now we have amplifiers U1 and U2 working, driving R9/R10 attenuator, and also driving our spectrometer through J2. Amplifiers U3 and U4 have their inputs grounded with capacitors. The A3038X and spectrometer are inside a Faraday enclosure, and we have a 10-dB attenuator between J2 and the spectrometer. We see P = 2.1 V and B has peaks 1053 MHz −10 dBm, 886 MHz −50 dBm, and 929 MHz −55 dBm. We replace L2 with a wire link. We see P = 1.9 V. The spectrum of B contains peaks 1226 MHz −11 dBm, 1297 MHz −22 dBm and several others. Remove R1 and see P = 1.6 V and B has peaks 1220 MHz −21 dBm, 1261 MHz −26 dBm. Restore L2 and remove J1. See P = 2.1 V and B has peak 1061 MHz −4 dBm. Remove U1 and replace with wire link, P = 0.8 V, B has peak 1217 MHz −42 dBm. Replace L1, P = 2.1 V, B has peak 1037 MHz −7 dBm. Load 27 Ω for R1, R2, and R17. Now P = 0.8 V, B has peak 1221 MHz −42 dBm. Connect test transmitter, transmit off, peak 929 MHz −44 dBm. Turn on −38 dBm transmit signal and see P rising to 2.2 V. We load R1 = R2 = 15 Ω and R17 = 75 Ω. With J1 open circuit, B has peak 1010 MHz −10 dBm. With J1 connected to 50-Ω coax peak is 929 MHz −52 dBm and B is 0.8 V. With L1 loaded and J1 open, P = 1.8 V and B has peak 961 MHz −17 dBm. We load UPC2746T in place of U1, and restore R1, R2, and R17. With J1 open and L1 loaded, B has peak 840 MHz −12 dBm. Connect J1 and peak vanishes. Restore BGA2803 for U1 and load 0 Ω for R1/R2 and 50 Ω for R17. On P 2.1 V, on B 1012 MHz −8 dBm. Remove R1 and see P = 0.9 V. Remove R2, P = 0.9 V. Load R1 = R2 = R17 = 51 Ω. Have P = 1.0 V, B peak 929 MHz −46 dBm. Connect −68 dBm SCT signal and see P rise to 1.2 V during burst.
With L1 and J1 loaded, and R1 = R2 = R17 = 51 Ω, apply −8 dBm SCT signal to A3015C loop antenna. A3038X in Faraday enclosure. Can see signal clearly on B at range 50 cm. Take A3038X out of enclosure and hook up to P and D as well as trigger from SCT. Apply −48 dBm SCT to J1. See SCT signal clearly on DA, and P rises from interference level 1.5 V to burst level of 2.0 V. Disconnect SCT and connect spectrometer to B, see −27 dBm in 902-928 MHz, with P varying 1.0-1.5 V. Connect −8 dBm SCT signal to A3015C. Reception range outside Faraday enclosure is only 10 cm.
[16-SEP-20] We restore R1 = R2 = 8.2 Ω and R17 = 150 Ω. We remove L1. We see P = 2.22 V as the circuit oscillates. We have 47 pF capacitors in P0603 package (Digikey 445-1277-1) with self-resonant frequency 900 MHz. We connect directly from pin U1-2 to C3-2 to add local decoupling. Now P = 1.24 V. We add 47 pF to U2 in the same way, P = 1.16 V. We load a second 47 pF in parallel to the one next to U1 and P increases to 1.64 V. So we remove that capacitor and add one to U3 and U4, so each has its own local 47 pF. Now P = 1.10 V. Again we double-up 47 pF next to U1, again P increases to 1.64 V. We move the two 47 pF and mount them on C3, P = 2.02 V. Remove one 47 pF, leaving one 47 pF on C3, P = 2.08 V. Replace C3 = 100 pF and the 47 pF addition with a single 47 pF, P = 2.04 V. Restore 47 pF next to U1, and load 47 pF for C3, see P = 1.84 V, which turns out to be −21 dBm of 1200 MHz. Replace C3 with 1.0 nF, P = 1.16 V. We remove R1 and P = 1.02 V. We replace C3, C5, C8, and C11 with 1.0 nF and each of U1-U4 has 47 pF soldered to pin two. We see P = 1.04 V. Now we double up the 47 pF on U1, and still see P = 1.04 V. So we remove the two 47 pF and stil see P = 1.04 V. Remove all 47 pF and P = 1.12 V. We restore the 47 pF and see 1.04 V again.
We have R1 = R2 = 8.2 Ω and R17 = 150 Ω, no L1, and combined 47 pF and 1.0 nF decoupling capacitors. We connect −60 dBm of 910 MHz through a 1-m cable to J1 and see −20 dBm at J2, suggesting gain at least 40 dB. At J3 we see +0 dBm and at J4 +2 dBm. Apply −40 dBm input and see +0 dBm at J3 and +1 dBm at J4. Outputs J3 and J4 are after −3 dB attenuators, and we are adding an additional 50-Ω load when we make our measurement, so saturated output power of U3 and U4 appear to be +3 dBm and +4 dBm. According to the BGA2803 data sheet, saturated output power should be −3 dBm. We apply SCT signal to J1 and find we need at least −38 dBm to get demodulated signal at D, even in Faraday enclosure, even when waiting for interference to subside.
[17-SEP-20] We go back to decoupling with 100 pF on their footprints and restore R1 = R2 = R17 = 51 Ω. We apply −58 dBm SCT signal to J1 and see clear demodulated levels on D. We apply −60 dBm from our synthesizer and measure power at B, J3, and J4 by plugging our spectrometer into J2, J3, and J4. We do not remove any resistors. When we plug our cable into J2, J3, or J4, the signal at the connector is loaded by two 50-Ω impedances in parallel. We measure power at each frequency by finding the peak in our spectrometer and observe the following ±5 dB variation in gain in the SAW filter pass-band. (NOTE: On 24-SEP-20 we obtain ±2 dBm gain uniformity within the SAW passband after exchanging L2.)
[18-SEP-20] We have R1 = R2 = R17 = 51 Ω, and no L1. Now we load in parallel with the C3, C5, C8, and C11 10 nF and 47 pF, with the 47 pF right next to the amplifier pins. We have no L1. With J1 open-circuit, P = 0.58 V. We detect −58 dBm SCT clearly on D when 929 MHz interference we pick up with spectrometer antenna is −60 dBm. With −47 dBm interference, D is contaminated with what looks like 50 MHz noise. We apply −47 dBm sweep to J1 and observe D and P.
When we drop the sweep to −56 dBm, we must wait until 929 MHz interference dies down to −69 dBm before taking our photograph.
We load L1 and place A3038X in Faraday enclosure. We look at D while moving a transmitter from one place to another. With the door closed, we see the transmit burst in all locations, and the data bits are clear for ranges up to 10 cm. When the transmitter is farther away, the bits are overwhelmed by 50 MHz noise. We note that the A3027E's superhet receiver provides a total gain of 100 dB with limiting at 6 dBm (±0.7 V) before demodulation, while this amplifier provides 80 dB gain before limiting at −5 dBm at J4.
[21-SEP-20] We have R1 = R2 = R17 = 51 Ω, L1 loaded, apply −40 dBm of 915 MHz to J1. See P = 2.12 V. Load 0.5 pF parallel with L1, see 2.12 V. Load 1.0 pF see 1.40 V, load 1.5 pF see 1.32 V. Remove parallel capacitance.
We have R1 = R2 = 8.2 Ω, R17 = 150 Ω, L1 loaded. Apply −40 dBm 915 MHz, P = 2.36 V (−5 dBm). Apply −50 dB=m, see 2.12 V (−13 dBm). We remove L1. We remove R20, R21, DA1, and DA2. We replace R9, R10, R12, R13 with 0 Ω. We apply − 60 dBm of 915 MHz and see −5 dBm on J4. We remove R19 and replace R6 and R7 with 0 Ω. Apply −50 dBm to J1 and see P = 2.16 V (−12 dBm). Restore R19, R6, R7 and see P = 2.08 V. Removing our 3-dB attenuator between L2 and U2 gives us a 2-dB increase in gain. Of our six attenuators, we leave the first three in place to stabilize the antenna and filter. We remove the fourth and fifth to increase the overall gain. We leave the sixth one in place because it makes no difference to the relative power of any signal, and offers better isolation of the switched capacitor filter. We apply −58 dBm SCT signal to J1 with L1 loaded. When interference dies down we have a clear 200 mVpp SCT signal on D = 4R. With −64 dBm we never get a clear signal.
[23-SEP-20] We measure the reflection coefficient of J1 with L1 loaded and R1 = R2 = 8.2 Ω, R17 = 150 Ω. We apply −20 dBm to the OUT terminal of a ZFDC-10-5+ directional coupler. We connect CPL to our spectrometer 50-Ω input. We connect IN to a 1-m coaxial cable. With 50-Ω termination we see −52 dBm at CPL, suggesting reflection of no more than −41 dBm, or 1% of incident power. We remove terminator and see −29 dBm, suggesting −18 dBm reflected. We call that 100%. now connect to J1 and see −48 dBm on CPL, suggesting −37 dBm reflected power, or roughly 2% reflection.
We remove L2, the SAW filter, so we can see the response of the demodulation tuner. We apply a −26 dBm sweep.
We adjust C16 and C17. We have L3 = 3.3 nH. We measure the frequency and height of the peak in D
We restore C16 = 0.5 pF, C17 = 2.0 pF and restore L2, the SAW filter. We apply a −26 dBm sweep.
[24-SEP-20] We apply 4.5 V power supply to our A3038X by mistake for twenty seconds, but the board seems fine afterwards. Current consumption 33 mA at 3.0 V supply. We replace L2, the SAW filter. We remove L1. We use J2 to bring B to a ZAD-11 RF input, and mix with 820 MHz LO. We choose our LO frequency so it is well outside the pass-band of the SAW filter. We obtain a clean sweep on D for 1000 MHz LO as well. We view the IF on the scope, as well as D and P.
We load L1 and connect a −3 dBm sweep to an A3015C loop antenna. We place the antenna 20 cm from L1. We wait for interference to die down and take the following picture.
We remove L1 and apply −48 dBm SCT signal to J1. When interference subsides, we get clear SCT logic levels on D.
[23-DEC-20] We have A303801A boards for the A3038DM-A RF Detector assembly, schematic S3038A_1 and S3038A_2. We assemble first amplifier stage with R1, R2, R3, and L1. We generate 910 MHz with our synthesizer, carry it through 1.5 m of cable to P1 (antenna input A) and then extract the output from P2 (filtered output B). We do not load the antenna inductor, L1. We have no other load downstream of P2. We forget to include the 10-nF and 10-μF decoupling capacitors on the 3.3-V power supply, but the amplifier performs as below. Current consumption is 6.7 mA.
We add the decoupling capacitors C5, C6, C7, C22, C23, C24, C25. We set the synthesizer to −20 dBm and vary frequency, then repeat with −30 dBm.
In the pass-band of L2 B3588 with −20 dBm input, we see −10 dBm output at B. Assuming 1 dB loss in delivery cables and connectors, 3 dB insertion loss for L2, and 3 dB attenuation in each of R1, R2, and R3, gain in U1 BGA2803 is +24 dBm, as specified by the data sheet.
[24-DEC-20] We connect an A3014MT −3 dB output to A through 20-dB attenuator and 1.5-m cable. We measure −17 dBm at B, 916 MHz. We add U2 LT5534 and C31. Current consumption 14 mA. We terminate B with 50 Ω. We sweep A from 850-950 MHz and look at P.
The detector output ranges from 0.2 V to 2.2 V. According to the data sheet, this corresponds to −60 dBm to −5 dBm. We touch the amplifier circuit and the response at P remains the same, except for an overall drop of about 0.2 V when we enclose the entire signal chain in our hand. We load U5, U6, U7 BGA2803 with coupling capacitors. Current consumption is now 36 mA with no input, 39 mA with −10 dBm on input. We have no R5, and connect hand-held spectrometer to P3 (C) via 10-dB attenuator. We apply power to A with synthesizer through attenuators as needed and 1.5-m cable. We measure power at C versus power at A and add to plot above. Power at C increases by 20 dB as we increase A from −70 dBm to −60 dB, which suggests domination of our input by interference.
We load R5, a solder lump for C14, detector diode D1 SMS7630, and 100 pF for C16. Schematic has D1-1 and D1-2 swapped around, we rotate D1. We take the signal from P5 (C) to a ZAD-11 mixer through −10 dB, mix with 915 MHz, and display on scope along with P and R. For A we have a −33 dBm sweep.
The IF is 30 mVpp or −17 dBm. Loss in the ZAD-11 mixer is 7 dB, so C is 0 dBm. We are sharing C between P3 and R5, so C without extraction at P3 would be around 3 dBm, consistent with our plot above. At the output of R5 we should have −3 dBm. Our calibration of the SMS7360 leads us to expect a 100-mV rectified voltage R. We see 20 mV, and outside the pass-band of L2 we see 60 mV. We rotate the diode, switch the diode, change capacitors, isolate the diode output, and no matter what we do, the output is never what we expect. The diode itself has marking XD8, which is correct for SMS7630-006LF.
[26-DEC-20] We build a power detector out of a 50-Ω resistor, 100-pF capacitor, and SMS7630 diode, soldered to the end of a short coaxial cable that we can plug into P2 or P3. We apply −33 dBm 915 MHz to P1. We get 100 mV on our detector when plugged into P3 with R5 removed. Our spectrometer says −1.5 dBm of 915 MHz. If we disconnect the 915 MHz input, we get 200 mV at our detector, implying +1.5 dBm wide-band power at C. If we apply a sweep, we see a 100 mV in the pass-pand of L2, but 200 mV outside the passband. With no input on A, we scan the power at P3 (C) from 250-6100 MHz and see a dozen peaks as high as −29 dBm, but nothing higher. The power at C is greater if we are amplifying interference than if we are amplifying a carrier frequency.
We build the tuner with R5 = −3 dB, C14 = 0.5 pF, C15 = 3.2 pF, L3 = 3.3 nH. We have D1 oriented correctly. We load 100 pF for C16 and omit R6. We apply our −33 dBm sweep again and look at R and P.
The peak of the tuner response is at around 945 MHz, where the attenuation by the SAW filter (L2) is around 30 dB. The −33 dBm sweep at 945 MHz produces −60 dBm at B. The gain of U5-U7 we expect to be around 72 dB, so the output of U7 (C) will still be saturated at 945 MHz.
[28-DEC-20] We load C16 = 10 pF, R6 = 1 kΩ and complete the circuit shown on S3038A_1, ending with the production of logic signal Q at U9-6. Current consumption is 40 mA for 3.3-V power supply. We try various combinations of capacitors and inductors in the tuner, settling upon C14 = 1.0 pF made out of two 0.5 pF in parallel, C15 = 3.2 pF made out of 2.2 pF with two 0.5 pF in parallel, L3 = 3.3 nH. We connect a −58 dBm sweep to P1 (A) and look at P, R, and the RF input mixed with 915 MHz.
We apply −34 dBm SCT signal to A and look at P, R, and F, this last being the output of the demodulation band-pass filter. We load 1.0 kΩ for R6, but we drop C16 to 2.2 pF from our original 10 pF as this improves the fall time of R.
We compare the SCT bit sequence to our receiver output, Q, which is supposed to be the recovered SCT bit sequence for our −34 dBm SCT input.
The received signal is always the same as the original. We drop the SCT power to −44 dBm and the received signal is the same 90% of the time, but we are seeing interference corrupting reception. We are operating out in the open on our bench.
[31-DEC-20] Load remaining components onto A3038DM-A and program U10 with prototype message detector P3038A_Main. We do not put U10 into standby mode. Current consumption of entire board is 47 mA.
[01-JAN-21] We assign LED4 (white) as a message receive indicator and LED5 (blue) as a message ready indicator. The former illuminates when we are receiving a message, but the message has not yet been decoded and accepted. The latter illuminates only when the message has been accepted. We use LED1 for power and we flash LED2 and LED3 to show clock function. Our message decoder is a VHDL translation of the original ABEL decoder P302702A12_Decoder used in the Octal Data Receiver. We connect test transmitter −44 dBm signal to A with P1 and get 100% reception. We remove P1 and load L1. We get 100% reception from an A3028E at ranges 0-2 cm, and some reception at ranges up to 10 cm. In a Faraday enclosure, we have robust reception up to 4 cm from L1, but no farther. Current consumption is 52 mA.
[04-JAN-20] We note that power measured at P is 10 dB lower than we expect. We replace L2, the SAW, and power returns to our previously-measured values. We replace C16 with 10 pF and reception of SCT signal improves. We attach a 3.3-V regulator to a lipo battery and supply the board with clean power. We see a reduction of noise on D. We connect −48 dBm SCT signal to A and obtain intermittent reception. We connect an A3015C loop antenna to A and place in Faraday enclosure with an A3028E. We obtain robust reception at ranges up to 10 cm. We load a 230-nH inductor antenna for L1 and obtain no reception. We are fiddling around and replace U1 and L2 again, but cannot obtain reception from −38 dBm SCT signal. We have 10-MHz noise on D.
[05-JAN-21] We reflow capacitor solder joints, replace connector P1, and clean. We examine the response to an SCT sweep and note roughly 10-MHz ripple in the step-response of D. We place 10-nF capacitors in parallel with decoupling capacitors C3, C8, C9, and C10. The ripple in D persists. We connect our A3014MT modulating transmitter to A through −20 dB and drive it with a 200-mVpp 1-MHz square wave to view the step response of the tuner and detector circuit. We look at D, which is R amplified by five. The signal applied to A is −24 dB of 900-920 MHz.
[06-JAN-21] We remove 10 nF capacitors in parallel with C3, C8, C9, and C10. We remove the tuner and connect R5 directly to D1. We apply RF to A with our synthesizer and find oscillations in D of several Megahertz for constant frequency and power applied to A. As we vary the input power and frequency, the amplitude and frequency of the oscillations varies. We replace C1, C2, C32, C11, C12, and C13 with 10 pF. Previously, when loaded with 100 pF, the corner-frequency of the high-pass filter made by the coupling capacitor and the 50-Ω input impedance of the next amplifier stage, was 32 MHz. Now the corner frequency is 320 MHz. The oscillations of several Megahertz decrease in amplitude by a factor of two.
We apply a −54 dBm sweep and find D to be cleaner than before. Note that we are powering the circuit with the LiPo battery and regulator.
We apply a −24 dBm sweep and observe oscillations on the demodulator output are reduced by a factor of two. We still see the spikes at around 400 kHz, but they are smaller. Note that these spikes exist despite our using a LiPo battery as a power supply, so they are not caused by our bench-top power.
We connect an A3015C loop antenna to A through P1. We place in Faraday enclosure with an A3028E. We obtain reception from 90% of locations within the enclosure, at ranges up to 60 cm across the diagonal. Reception at less than 10 cm is robust.
[07-JAN-21] We apply −60 to −10 dBm 927 MHz to antenna input A with our synthesizer and watch demodulator output D. We see oscillations of 7.5 MHz on D and amplitude 40 mV from −50 to −10 dBm. At lower power, D becomes more chaotic, but we still see the oscillation buried in the noise. We vary the frequency and keep power at −30 dBm.
The oscillation is consistent with an intermediate frequency (IF) created by mixing the amplified synthesizer frequency at C with a fixed interference frequency of 920 MHz picked up by the split-capacitor tuner C14-C15-L3. The amplified synthesizer frequency on C acts as the local oscillator (LO) for mixing at the detector diode D1, and the 920 MHz interference acts as the radio-frequency (RF) input to the mixer. Diode D1's non-linear response to the sum of the LO and RF creates the IF at R. We note that we have a 40-MHz 3.3-V oscillator, U11, on the board, and it's 23rd harmonic will be exactly 920 MHz with amplitude 170 mVpp. This oscillator is mounted directly below the detector diode D1, on the other side of the ground and power planes. So long as the signal applied to A is −60 dBm or higher, we will have LO of 0 dBm, generating roughly the same amplitude RF. The IF amplitude is roughly constant, while its frequency is exactly equal to the difference between the synthesizer frequency and the 23rd harmonic of the oscillator. The 920 MHz RF signal is amplified by the split-capacitor tuner as shown above. When the LO is 915 MHz, right in the middle of our 902-928 MHz ISM band, the IF is 5 MHz, which is the bit rate used by our subcutaneous transmitters (SCTs). The IF passes through the discriminator (C17-19, R9-10) before arriving at the comparator as F. The frequency response of the discriminator is the A3007D trace in this plot. We have maximum discriminator gain at 5 MHz. The 920 MHz interference is optimal for disrupting SCT message reception.
We will try removing U11 to see if the oscillations disappear. We can try 80 MHz for U11, for which the 11th harmonic is 880 MHz. Its amplitude will be roughly double that of the 23rd harmonic of 40 MHz, but the gain of the tuner at 880 MHz is only a quarter of the gain at 920 MHz, as shown above. The IF frequency for 915 MHz mixed with 880 MHz will be 35 MHz, for which the gain of the discriminator is a third of its gain at 5 MHz. So the oscillation for LO of 915 MHz will be 2 × 0.25 × 0.33 = 0.17 the amplitude we see now, or six times smaller.
[08-JAN-21] We remove U11 and the oscillations disappear. We load a new U11 and the oscillations are back, same amplitude and frequency as before. We add 100 pF in parallel with C25, no change in amplitude. Remove both capacitors, no change. Restore 10 nF for C25, same as ever. We order 80-MHz oscillators.
[11-JAN-20] In Faraday enclosure, with 250 nH inductor for L1, battery power, we obtain robust reception at up to 5 cm from an SCT.
[12-JAN-20] We replace U11 with 80 MHz and apply −30 dBm to A. We observe oscillation on D of 20 mVpp, 12 MHz for input frequency 892 MHz, which is consistent with the eleventh harmonic of 80 MHz mixed with 892 MHz. At 915 MHz we see 10 mVpp, no oscillation. We reprogram the A3038DM-A to divide 80 MHz down to 40 MHz. We obtain robust reception with −38 dBm SCT signal and intermittent reception with −48 dBm SCT signal both outside Faraday enclosure. Use battery power inside enclosure and see intermittent reception of −58 dBm. Remove test SCT signal and move SCT around in enclosure, obtain robust reception at ranges up to 10 cm and intermittent reception at ranges up to 50 cm.
[03-FEB-21] We receive A303801B printed circuit boards for the A3038DM-A Detector Module. They have one error on the silk screen: they are marked A303801A.
[02-MAR-21] Working with an A3038DM-A, we program the readout of the ADC to be made every time we get a RECEIVE signal assertion, and then illuminate LED2 and LED3 using the power level. We see the power level rising and falling so long as we get preliminary reception.
[09-MAR-21] We have two A3038DM-B. With C14 = 1.0 pF and C15 = 2.2 + 1.0 = 3.2 pF their tuner resonates at around 935 MHz, giving a significant positive slope through the SAW filter pass-band. We do not load L1. We remove L1 from our A3038DM-A. We apply −58 dBm sweep to
[10-MAR-21] We apply a sweep to A for each of our detector modules. We have what we believe is −8 dBm followed by attenuators. We look at P and see once again with −58 dBm sweep and see that the peak is 0.2 V higher in No0 than No1 and No2. But we also note that the value of P outside the sweep is also 0.2 V higher in No0 than in No1 and No2. Looking at the LT5534 data sheet, the output with no RF signal has range 0-380 mV. The No0 detector appears to be at the high end of this range, at 0.4 V, while the No1 and No2 detectors are in the middle at 0.2 V. We look at D and first see signs of failure of the demodulation ramp at −61 dBn for No0 and No1, and at −58 dBm for No2. Reception with −38 dBm SCT signal is 100% for No0 and 0% for No1 and No2. It turns out that C16 on No1 and No2 boards is 10 pF. We drop to 2.2 pF. We see poor reception at −38 dBm. We increase the gain of the demodulator amplifier by dropping R7 from 270 Ω to 100 Ω so that the gain is now ×11. We obtain 95% reception down to −51 dBm with all three boards. We are using our bench-top power supply to provide 3.3 V to the A3038BB. When we switch to the PoE power supply, which contains an upside down converter, reception at −51 dBm drops to 10%, but we see robust reception at −48 dBm.
[11-MAR-21] We load 2014VS-111ME for L1 in No1 and No2. We do not connect the inductor to ground, but instead isolate it from the ground pad with a paper barrier. The inductor acts as an open-ended coil antenna. We power the A3038BB-X with a lipo battery and regulator. We enter an FE5A faraday canopy and move a transmitter around over the two detector modules. We see robust reception up to range 10 cm, and intermittent reception at 20 cm. The white LED is acting as a power indicator, and we see power variation dominated by range and little affected by orientation. We connect the No2 inductor to ground and move transmitter close to one then the other. Power and reception by the ungrounded inductor is steady and robust, while over the grounded inductor we see power and reception vanishing with the transmitter in unfavorable orientations close to the detector. We now suspect the A3032 tracker suffered from two serious problems. One was the shifting of the eight-bit ADC power value one place too far to the left, so that power 0x80 would come out as 0x00, causing the measured power to drop when the transmitter was close to the coil. Another was the grounded coil being sensitive to the polarization of the incoming microwave. The coil itself consists of 3.5 turns of diameter 10 mm, with 10 mm of extensions for mounting, making a total length of around 110 mm. The wavelength of 915 MHz is 330 mm, so our coil is a 1/3-wave antenna.
[12-MAR-21] We solder the dummy contact on L1 to its pad, cut the ground contact to create the coil antenna shown below.
In our Faraday canopy, with battery power supply, we obtain robust reception at ranges up to 12 cm from the detector coil, including the vertical direction. Between two neighboring coils, up to height 12 cm we obtain reception from at least one of the coils always, which is to say: reception is 100%. Detected power always decreases with range, regardless of orientation.
[22-MAR-21] We complete the eight-page A3038C circuit diagram for the ALT based board. We select a 24-way flex connector HLW24S-2C7LF in place of a non-existent sixteenth coil, so that we can contatinate ALT base boards into larger trackers.
[29-MAR-21] We have four more A3038DM-B without L1 loaded. We generate 820-980 MHz, −34 dBm sweep and apply to A. After correcting one joint, we have four perfect responses on P. We apply sweep to P2 and adjust split-capacitor tuners. We have C14 = 1.0 pF and C15 = 2.2 pF. We add 0.5 pF to C15 on two boards and see a fine ramp 900-930 MHz. We see this same perfect ramp all the way down to −54 dBm on the two new boards, but on the old board with L1 removed the ramp already has dips. We find that the four new boards have 10-MHz oscillators, so we set up a PLL in the logic chip to generate 40 MHz from 10 MHz. The result is 100% reception with −51 dBm SCT signal on both new boards, but only 80% with older board powered by 80-MHz oscillator. Current consumption of the new boards with the PLL is 59 mA, while older boards is 51 mA.
[02-APR-21] Having loaded L1 onto our four boards with 10 MHz oscillators, we find they pick up messages from interference, while our original 80-MHz boards do not. We replace 10 MHz oscillators with 80 MHz and re-program. Now we examine reception with six detector modules in faraday canopy with one transmitter and powered by PoE. We are able to position the transmitter 5 cm over the platform and receive the signal with none of the six. Some of the six are more sensitive than others. This result contrasts with our earlier robust reception with battery power and two detector modules.
[03-APR-21] We number all six of our A3038DM-B. No1 and No2 are the first two we made, which today are the only ones with test point wire loops loaded. We load four into the corners of the base board test fixture and two into the center two spots. We sit in our Faraday canopy with transmitter Q154.7 and the base board on the floor of the enclosure, insulated from the aluminum plate by a plastic-coated carbon sheet. We power the base board with a 3000 mAhr lipo battery and see No1 in a corner position provide 99% reception all over the platform. The other modules perform well, but not quite as well as No1. We switch to PoE and we see reception from No1 drop to 80%, with the loss a function of position and time, so that the receiving light flickers in an unfavorable position, and goes out occasionally in a favorable position. The other modules suffer the same time-scattered losses.
[05-APR-21] We add 3 μF of decoupling on 3V3 next to two corner detector module sockets. We apply PoE power. We believe we see some improvement in reception from all detectors. The two with decoupling do not stand out among the others. We compare tuning on our six modules. The No1, which performs best, has the following tuning response.
The peak response of No1 is at the top edge of the SAW pass-band, which is roughly 930 MHz. We look at No6, which performs less well than the others, and its peak respose is at around 960 MHz. We add 0.2 pF P0402 to C15 and f_peak is 940 MHz. We add another 0.1 pF and it's at 930 MHz. We continue through the other detectors, setting f_peak in the range 930-935 MHz. If we do not wash off the water-soluble flux residue after adding a capacitor, the top of the tuning response will be flattened and its frequency lowered by about 5 MHz. So we take care to wash and dry after soldering and re-measurement.
We load all six detectors onto our base board, power with LiPo battery in enclosure and move transmitter around in the entire enclosure, antenna in air. At least two of the detectors receive the signal from every location in the enclosure. All detectors attain 95% reception for the transmitter movingh around over the platform, even those in the corners. We apply −50 dBm sweep to B and see inadequate to adequate sweep response on D. If inadequate, raising the sweep power to −44 dBm gives an adequate sweep. We change the UMC connector we are using to deliver power and now all boards give adequate sweep with −50 dBm sweep applied to A with L1 loaded.